'jlh'에 해당되는 글 3건

  1. 2010.06.22 1969년식 JLH 의 온전한 구현
  2. 2010.06.22 여러가지의 JLH 버전들 (2)
  3. 2010.01.13 John Linsley Hood 의 Simple Class A Amplifier 번역(JLH AMP)
2010.06.22 02:32

1969년식 JLH 의 온전한 구현

오디오 DIY 맨으로 JLH를 몇개 만들었지만 아주 온전하다고는 볼 수 없었는데 
출력석을 내 마음대로 썼기 때문이다. 

얼마 전 김형섭님이 완전한 JLH 의 워킹카피를 만드셨는데 전원부까지 완전 복원된 셈이다. 
방열기가 작지만 케이스에 넣을 때 큰 것으로 넣으면 되니 별 문제는 없다. 

아주 좋은 소리를 내는 DIY 오디오의 중요한 앰프하나가 완전 복원된 셈이다. 
케이스에 들어가면 아주 좋을 것 같다는 생각이 든다. 
긴 세월동안 들을 수 있는 얼마 안되는 앰프중에 하나다, 거하지도 않으며 10와트 정도의 출력으로
잘 조정되면 소리는 정말 최고다. 클래스 A의 성배같은 앰프중 하나다, 
나머지는 알레프와 F5 , hiraga 같은 앰프다. 물론 크렐 ksa-50이나 다른 상업용 앰프들도 중요하다.
그러나 10와트 언더에서는 오리지널과 양전원 버전은 정말 최고다. 

많은 앰프를 들어보았어도 변함은 없다.

회로는 오리지널 그대로 .. 

많은 사람들이 만들어 보았으면 하는 앰프이기도 하다. 
벤치마크 앰프로 수많은  영감을 낳게한 회로로 40년 이상 사람들의 사랑을 받고 있다. 
어느 정도 많은 사람들이 만들어 보면 음질과 회로에 대해 토론이 될 것 같기도 하다. 



저작자 표시
신고
Trackback 1 Comment 0
2010.06.22 02:31

여러가지의 JLH 버전들

이상하게도 이 글은 비공개로 되어 있었다. 
별 내용은 아니지만 무언가 더 끄적거리고 공개로 바꿀 예정이었던 것 같다.

1.  DOZ를 제외한 버전으로 제작에 참고한 것은 Nob라는 일본 사람의 글이다.
半導体アンプの7台目です。

JLH Class-A Ampを作成してみました。
このampは1969年にJohn Linsley HoodさんがWireless World という雑誌の4月号に発表した回路を基にしています。
その後1996年にupdateが発表され、1996年versionとして海外では数多く作られているようです。
さらに2003年にupdateが発表されています。
今回はこの2003年版を基に作成しました。

Circuit

回路はこのようになりました。

Parts

  • 出力段には2SC5200を採用しました。(千石で300円と安かったので)
  • ドライバ(と言えるか)には2SC3421。(これも千石で50円と安かったので)
  • 定電流回路には2SA1538を。(2SC3421のペアでこれも50円)
  • 初段およびその他の定電流回路には2SA1015。(秋月で1000円/200個)
  • 抵抗は0.33Ω5W以外、在庫があるものは金属皮膜、なければカーボンです。
  • コンデンサですが、電解はふつーのもの。カップリングのC1は結局外しました。

Making

試作したところ、やはりA級です。発熱が半端ではありません。
放熱を考えて、左右を別ケースとして、ケース自体を放熱器とすることにしました。

Adjust

半固定抵抗のVR1,2,3は中点にしておきます。
電源ONにして出力電位をチェックします。
VR1で出力電位を0Vに調整します。調整しきれないときはVR3を調整してください。
R10(0.33)の両端の電圧を測定し、0.33で割った値が出力電流となります。
出力電流はVR2で調節します。オリジナルでは1.5Aとか流していますが、0.5Aで充分でした。
この場合0.33Ωの両端の電圧は0.15V程度です。

Impression

これはもう、作ってみるしかないです。
低音の響きの豊かさは、これまで聞いていたものはいったいなんだったの、、、状態です。
特にウッドベースの音にはしびれます。

2N3055 version

その後、秋月で1つ150円で売っていた2N3055を使って、もう1セット作ってみました。
10個購入し、hFEを測定してpairをとりました。
hFEは、かなりバラツキがあるんで、面倒でもpairをとったほうがよいです。
2N3055はTO-3型なので、後面にTO-3用のheat sinkを付けるようにしました。
上からみた配置です。すかすかです。
後ろから見た図です。中心が電源。右に入力、左に出力。下側に2N3055が実装されています。
基板表面。
基板裏面。

2.  그런데 이 앰프는 본질적으로 2003 업데이트의 일본판이다,

JLH Class-A Update

 

 

I had originally intended that this page would be a step-by-step record of the modifications carried out during the past year by one constructor – Tim Andrew. However, recent ill health has meant that I have been unable to spend much time sitting at my pc so, rather than incur yet more delay in publishing the results, I have decided to write a short summary instead. I am very pleased that Tim has taken the time to supplement this with his own comments. At the end of the page is a brief update on the higher power ‘JLH for ESL’ circuit.

 

Tim is a professional musician (a classical concert pianist) and so I trust his subjective judgement when it comes to assessing the accuracy and realism of sound reproduction. Before Tim first contacted me, he had built a kit version of the 1996 design, which he had subsequently upgraded with higher quality components. Though Tim was happy with the results, he was keen to see if further improvements could be made to the sound quality and I was pleased to be able to suggest various circuit modifications, the majority of which subsequently proved to be very worthwhile. Each of the modifications was carried out separately so that the results could be evaluated on an individual basis.

 

Rather than show schematics for each stage, I will start off with the penultimate circuit and include some appropriate comments.

 

 

Fig 1 – The Penultimate Circuit

 

Transistor substitutions

 

One of the first modifications was to try alternative output transistors. The MJL3281A gave an audible indication of oscillation and was quickly rejected. The MJ21194 sounded significantly better than the 2N3055 but, in Tim’s layout, introduced a low-level hum. The MJ15003 gave a similar improvement to the MJ21194, but without the hum, and so was retained for future use. At a later stage, the BC212 and 2N1711 (Q4 and Q3) were replaced with the 2SA970 and 2SC3421.

 

Output dc offset control

 

The standard dc offset control circuitry (7815 and associated components) was replaced with a two transistor constant current source (Q5/Q6). I had various reasons for suggesting this change. Firstly, three terminal regulators are not renown for their quietness and so it did not seem like a good idea to inject the noisy output from one directly into the feedback loop. Also, I had received reports that certain 7815s oscillated due to the low current conditions under which they were being operated.

 

However, one of the main benefits of the ccs is that the output dc offset variation as the amp warms up is greatly reduced. This is because the temperature coefficient of the ccs acts in the opposite direction to that of the input transistor (Q4) and negates the effect of temperature changes in Q4 (assuming that the temperature of Q5 follows that of Q4). This cancellation of temperature coefficient effects can be put to further good use as will be seen later.

 

Quiescent current control

 

I first suggested that Tim try the 1969 bootstrap Iq control circuit, partly because the simulated distortion figures were half those for the 1996 version but mainly because I wanted to know how the two methods of Iq control compared in the same amplifier. I had received reports that the 1969 circuit (modified to dual supply rails) sounded better than the 1996 version, but I could not be sure that there were no other variables involved. As it turned out, the bootstrap circuit was a retrograde step and Tim immediately reverted to the original 1996 arrangement.

 

I still had some nagging doubts about the 1996 Iq control circuit and so I suggested introducing another constant current source (Q7/Q8). As with the bootstrap circuit, the simulated distortion figures were still half those for the 1996 version but with the added advantage that the distortion did not increase at low frequencies due to a reduction in capacitor effectiveness. A further advantage was an increase in amplifier efficiency (or maximum output). The maximum output voltage swing with the ccs is greater than that for the standard 1996 circuit and the maximum output current increases from around 1.35 to about 1.5 times the quiescent current.

 

When carrying out this modification, Tim reused the existing MJE371 for Q8. R10 has been retained to provide an easy means of measuring the quiescent current. To my relief, Tim found the second ccs to be worthwhile improvement.

 

Power supply

 

Whilst making the other alterations, Tim also took the opportunity to upgrade his power supply, initially by fitting larger bridge rectifiers and snubber capacitors and then by replacing the LM338s with ‘follower’ type discrete regulators, in line with my desire to remove unnecessary feedback loops from the overall circuit. The ‘follower’ regulators, basically a capacitance multiplier circuit with a fixed voltage reference (derived from a resistor fed by a ccs), gave a small improvement. A much greater improvement was obtained when separate regulators were provided for each amplifier, whilst retaining a common transformer, rectifier bridges and reservoir capacitors.

 

 

Fig 2 – The Final Circuit

 

Removal of the feedback capacitor

 

I had received emails from a couple of constructors reporting on the beneficial effects of removing the feedback capacitor (C4). I passed these comments on to Tim and he decided to try this modification for himself.

 

This modification should be treated with caution. I would not recommend trying it unless the dc offset ccs (Q5/Q6) modification has been done first because otherwise the output dc offset variation during the warm-up period is likely to be in the order of several hundred millivolts. In Tim’s case, with the dc offset ccs fitted, the output dc offset variation with the feedback capacitor removed was only slightly higher than that which he had previously with the standard 1996 circuit.

 

I believed that the offset variation could be reduced further by utilising the temperature coefficient of the Q5/Q6 ccs. I therefore suggested that R11 be made adjustable so that the temperature rise of Q5 could be varied. In this way, the output dc offset variation due to temperature changes in all stages of the amplifier could be compensated for, though this requires a lengthy, iterative process. With the amp at its normal operating temperature, the offset is adjusted to near zero using VR1. The offset when the amp is cold is then measured. VR3 is adjusted slightly, the amp is allowed to warm up and the offset is re-zeroed using VR1. The offset is then rechecked when the amp is cold and the process repeated until the minimum offset variation has been obtained. Tim has been able to achieve an output dc offset variation between switch-on and normal operating temperature of less than 50mV.

 


 

15/03/2003 Addendum

 

It has been brought to my attention (thanks Mietek and Rudy) that removing the feedback capacitor increases the hum level at the amplifier output, which is particularly noticeable with high sensitivity speakers and if a simple rectifier/capacitor power supply is used. I had not anticipated this, but some quick simulations soon indicated that removal of the feedback capacitor reduces the PSRR of the amp by a factor of about 3, causing any supply rail ripple to become more audible.

 

Fortunately, the cure for this problem is relatively simple. The PSRR of the input stage ccs can be improved by the addition of a single capacitor, connected between the junction of VR3/R11 (Fig 2) and the +ve supply rail. Doug Self’s ‘Audio Power Amplifier Design Handbook’ indicates that this modification will improve the PSRR of the ccs by about 10dB. A capacitor value of 47uF will suffice, but higher values (within reason) can be used.

 

The higher power (‘JLH for ESL’) circuit can be similarly modified by splitting R11 (Fig 3) into two 4k7 resistors in series and connecting the capacitor from the mid-point of these resistors to the +ve supply rail.

 

This modification can also be carried out even if the feedback capacitor is not removed, and will give an improvement in PSRR with the corresponding reduction in hum.

 

 


 

17/08/2003 Addendum

 

Several constructors have found that adding the 47uF capacitor to the input stage ccs after having removed the dc blocking capacitor from the feedback network has caused the ccs to become unstable. This has manifest itself by relatively large output dc offset variations when taking voltage readings around the input circuit or when a hand is moved near to the ccs components.

 

In Tim’s case, a successful solution to this problem has been to replace Q5 and Q6 with ‘slower’ transistors. The MPSA56 appears to work well in the ccs. Alternatively, the 47uF capacitor could be removed and the PSRR of the ccs improved by omitting VR3 and replacing R11 with a 1mA constant current diode (or an FET wired as a ccs to give a similar current).

 

Adding base resistors (100R to 1k) to Q5 and Q6 and/or a 1k resistor between Q6c and Q4e should also help to improve stability.

 


 

Tim’s comments on the modifications (Updated 17/08/2003)

 

A few years ago I built the 1996 version JLH Class-A amplifier. Constructors of this amplifier have commented about its smooth sound, with many favourable comments and comparisons against valve designs and a few not so favourable comments with regard to its limited power output. In its standard 1996 form, which I built from a kit using cheap components, my first impressions of its sound were of smoothness coupled with a relaxed liquid musical flow which I found far preferable to anything else which I had previously heard. In the context of my system with speaker efficiency somewhere around 87dB/W and with volume set correctly such as is appropriate for the perspective as recorded, or in other words "at a realistic level", its limited power output has never been a problem. The amplifier and its power supply have since been subject to extensive component substitutions and substantial circuit modifications.

 

As this section is about my impressions of the modifications that have been made to the circuit, a brief word on what I consider to be an "improvement" might be in order. I want to hear, with ease, the ambient signature of the recording venue, with a distinct impression of the space between its walls. Also, I want to notice, for example, the sound of the felt hammer of a piano hit the string, followed not only by the sound of the string vibrating but also the more subtle reflected and attenuated sounds of the hammer and its mechanism as these reverberate between the walls of the recording venue. This is sometimes more noticeable in larger venues where the reflected sound arrives later, albeit weaker. Those delicate piano harmonics must be reproduced with the greatest accuracy, enabling subtle shadings of timbre to be noticed, again with ease. As a pianist, I want to hear the "pitch" of the note as it decays through to its quietest moment as acutely as possible, but I want no hint of hardness or roughness. With orchestral strings for example, where there are many instruments playing together, I don't want to hear one homogeneous group, and I want transparency, not brightness.

 

Professionally, I have a very close affinity with the piano. A difficult instrument to reproduce, it is perhaps more revealing of faults in the reproduction chain than can be the case with other instruments although the human voice is also very useful, for obvious reasons. It is my view that any modification that produces a more realistic rendition of the complex sound of this instrument, and the very subtle structure of its over-tones, will also represent an improvement in the accuracy of the amplifier overall. This has been the case during all my listening trials. It is worth mentioning that any modification which leads to an apparent decrease, for example in the level of the treble, will not necessarily be deemed to be an improvement, even if the new treble level is a welcome one, unless it is accompanied by an improvement elsewhere, improved detail or portrayal of nuance for example. From this, you will gather that I am not in the habit of 'voicing' the system, adjusting one thing to correct for another, but that I prefer to address the transparency of the system as a whole, with the aim of neutrality. Only then will I look at altering the balance, perhaps with a slight adjustment to the treble. It is through this approach (transparency first, followed by tonal balance) that I am now able to enjoy the vast majority of recordings in my collection, previously I had found many of these to be deficient in one way or another. Almost without exception, each modification has improved "difficult" recordings, whilst further improving others, often revealing a warmth and atmosphere, the previous lack of which had been wrongly attributed to the recording.

 

Though considerable time has been expended on both the amplifier and its power supply, I find it sobering to say the least that improvements made to power supply, specifically to the method of its delivery into various parts of the amplifier circuit have been so rewarding. The following is a list of the modifications that, with considerable help from Geoff, I have been able to carry out on the 1996 version of the JLH. Also included are my opinions of the results of these. Each substitution has been carried out individually, this has enabled subsequent and hopefully accurate (but not always positive!) evaluation. !

 

The Amplifier

 

Input capacitor.

The cheap polycarbonate(?) 1uF input capacitor was replaced with a  470nF Mcap "Audiophile" polypropylene type.  This led to an improvement in both bass firmness and in detail, treble sounded less bright. Later, I replaced the Mcap 470nF with Audio Note paper-in-oil 470nF. This sounds very different, smooth, warm and open with much more textural detail and firmness in the bass. There is some loss of focus when compared with the better plastic types and the positioning of instruments within the stage is not as precise as it could be, however none of the plastic types I have tried has approached the naturalness and openness of the paper-in-oil, particularly in the treble, and any shortcomings are easily forgiven in light of considerable improvements elsewhere.  This simple modification has since proved to be one of the most effective. I have also tried a polystyrene type (333nF) which sounds more detailed and focussed than anything else tried previously, though there is a tendency to sound a little "squeaky" on occasions (placing a small paper-in-oil capacitor across it improves this considerably), nevertheless I prefer this to most polypropylene types, many of which sound hard and slightly blurred to me. 

 

Resistors.

All standard grade metal film resistors in both critical and semi-critical parts of the circuit were replaced with tantalum film types.

Improved smoothness and texture, with a more fluid sound. A slight "mumbling" quality has been removed.

 

Output transistors.

The 2N3055s were replaced with MJ21194. In comparison with these the 2N3055s sound grey and rather diffused with less sense of authority, less detail and a more prominent treble quality. In contrast, the MJ21194s have a noticeably firmer sound with more ambience in the treble and greater detail. More natural generally. Reluctantly, they were removed from the circuit due to a faint hum which was not present with the 2N3055s.

Wanting to try something else, and now with the strong impression that the 2N3055s were less than ideal, I tried some MJ15003s.

This time, a substantial improvement over the 2N3055s. The MJ15003's bass is both tauter and more authoritative, with cleaner treble and greater textural detail.

 

DC offset control.

Replace 7815 with constant current source.

Result...Cleaner, smoother and weightier, with what can only be described as an organic flow. It was obviously all there before, but I suppose it was masked somewhat by the noise of the regulator. The volume can be increased further without sounding "loud".  A substantial improvement in all respects.

 

Iq control circuit.

The Iq control circuit was replaced with a bootstrap circuit (using an Elna "Silmic"). Less clarity was the result, with less tonal variety and focus, sounding more shut-in. The bootstrap simply doesn't sound as detailed. I assume this is due to the presence of the bootstrap capacitor connected to the signal path. Perhaps a Black Gate might improve things, but I suspect not enough to equal the MJE371 circuit which is more transparent, open, dynamic and uncoloured, the female voice sounds less "female" with the bootstrap circuit. It strengthens my theory that those who prefer the earlier version of the JLH do so because of the absence of the 7815 in the earlier circuit. I would go further and say that due to the absence of both a bootstrap capacitor, and an output capacitor, and with the ccs in place of the 7815, they might well prefer the 1996 version, all other things being equal.  My original Iq control circuit was very quickly re-instated!

 

It was not long until the original Iq control circuit was removed again, this time replaced with a constant current source and with better results this time. The initial reaction is to think that the treble detail and "air" have been diminished with a reduction of transparency. On prolonged listening things are rather different. There is actually more detail coming across, coupled with a growing sense of "rightness". Sounds are presented in a more natural light, gone is the spotlight effect with its admittedly pleasant but artificial treble detail. String harmonics are more balanced and proportioned with a sense that they now belong to the fundamental, part of the whole. The gaps between rapid piano notes are often missed by amplifiers, the JLH reproduces these well and they are even clearer now than before. Familiar recordings of woodwind and brass instruments sound remarkably smooth and natural. Differences in scale between smaller chamber music recordings and larger scale works are now more clearly conveyed. It is interesting to compare the sound of the Iq ccs circuit with that of the bootstrap which shared many of the attributes of the ccs but had a lumpy and coloured, slightly congested characteristic which I found unpleasant. Returning to the standard 1996 Iq circuit the next day was quite a relief, this time I have no plans return. I would miss the qualities that the Iq ccs circuit has brought to the amplifier. Final thought........Recommended for those who want to sit down for an evening of good music and a fine wine.

 

Feedback capacitor.

The 470uF Oscon (previously a very similar sounding 220uF Silmic) feedback capacitor was replaced with link (needing a small change in value to the DC offset ccs preset). The result of this change was a more open and natural treble with an increased sense of fluidity, depth and ease. Hot/cold offset variation are much greater without the feedback capacitor, in my circuit a variation of 150mV was observed (with the feedback capacitor it was around 65mV), this was reduced by controlling the current through the ccs in an effort to adjust the temperature compensation, but on a recent re-build of the circuit this arrangement proved ineffective and was subsequently removed.   

 

Driver transistor (2N1711).

This was replaced with a 2SC3421. As with the other transistor substitutions I have made in the JLH, the actual pitch of a note is more easily heard with the 2SC3421s. The same characteristics introduced by the Iq ccs circuit are still there but each single note now conveys more "meaning", more clearly defined in time. Timing, of course, is a musician’s greatest asset! The Iq ccs circuit introduced a smoother, rounder sound with a somewhat darker hue, the extra transparency and openness brought about by the 2SC3421s has lifted that slight darkness away whilst apparently retaining the smoothness and naturalness of the Iq ccs.

 

Input transistor.

The BC212 was replaced with 2SA970 with similar improvements to those noticed with the 2SC3421.

 

The Power supply.

 

Rectifier diodes.

Having tried snubber capacitors across the original "standard" diodes with no noticeable improvement, the originals (and snubbers) were replaced with schottky types. This seemed to be beneficial with more smoothness and an improved "woody" quality with woodwind.

 

Regulators.

The LM338K regulator circuit was replaced with a capacitance multiplier. The bass now conveys more authority and the amplifier sounds a little warmer, also with more detail. 

 

Dual regulators.

The single capacitance multiplier was replaced with a new (adapted) dual version allowing separate regulation for each channel. This warrants a detailed write-up so I shall list my observations in the order in which I noticed them and in descending order of their magnitude.


It is only now that I have heard the new dual power supply, that I can identify the sonic effects of the single supply. For the first, and most important observation, I shall use a single piano note as an illustration. With the single supply, when the note is struck there is an initial transient 'bump' as the hammer hits the string, followed by the decay, which starts after the initial 'bump' has subsided. With the dual supply, this initial transient is less 'loud' (better controlled?) and it carries more weight and meaning, this is followed by the decay which not only conveys better pitch, leading to more emotion and tunefulness, but the decay starts sooner, its first moments not masked by the apparently exaggerated impact of the hammer blow introduced by the single supply. Also, due to the increased definition, the note seems to decay more slowly, incidentally this is one of the more significant differences between a small grand piano, and a large 'concert' grand where, due to the increased string length of the larger instrument, its sustaining power is much greater. A single note can therefore be followed more easily from start to finish. The tonal signature and real colour of all instruments are now better conveyed.


There is also a significant improvement in the quality of the treble where there is greater transparency. For most of the time, it is less obvious than before, and smoother, but little details previously almost un-noticed are conveyed more clearly and with improved texture. This treble improvement was unexpected and is a constant pleasure!


The third improvement I have noticed is an improvement in the positioning of individual instruments. The perceived stage width is not obviously any wider than before, although I couldn't fault it before, on a good recording the stage width was almost limitless, on a bad recording it had definite limits. This hasn't changed, what has improved is the positioning of instruments within the limitations of the stage width imposed by the recording, with instruments on the edge of the stage more clearly conveyed in space with a better "floating" feel to the acoustic coupled with a more acute sense of the venue.

 

Filter capacitors.

Having previously bypassed the standard grade electrolytics with Elna "Silmic" 100uF with little, if any improvement, this time the original capacitors (30,000uF per rail) were replaced entirely with "Silmics" (18,000uF per rail).  A superb improvement in definition. The scale of which came as quite a surprise.

 

Conclusion.

I consider the JLH in its present form, to be a very special amplifier. Its ability to portray the acute sense of emotion and excitement contained in a fine performance, through its accuracy and with such grace, coupled with its ability to scale music's dynamic heights so convincingly, is rare. My most sincere thanks to Geoff who, through spending so much time helping others like me, has so far not had time to carry out these modifications for himself *.

 

* Unfortunately not the only reason - Geoff

 


 

Higher power circuit

 

The ‘JLH for ESL’ circuit, which can be used with conventional speakers as well as electrostatics, already has a ccs for dc offset adjustment but it would benefit from the other modifications outlined above. In particular, the use of a ccs for quiescent current adjustment obviates the need for a high power preset, which can sometimes be hard to find.

 

 

Fig 3 – The Higher Power Circuit

 

When used with conventional speakers, this circuit can deliver over 40W provided the supply rail voltage and quiescent current are selected to suit a specific load impedance. The supply rail voltage needs to be a couple of volts higher than the peak output voltage swing and the total quiescent current should be about 0.7 times the maximum output current. The power dissipated in each output transistor (supply rail voltage times half the quiescent current) should be limited to about 40 to 45W, assuming decent sized heatsinks are used (0.6 to 0.8degC/W per transistor).

 

The peak load voltage and current can be calculated from required power and the speaker’s impedance in the normal way using:

 

Vpk = sqrt(2*Pwr*Rload)  and  Ipk = sqrt(2*Pwr/Rload)

 

To allow for speaker impedance variations, I would suggest that current is calculated using ¾ of the speaker’s nominal impedance and voltage using 1½ times the nominal value. Of course, you are free to make your own assumptions about speaker impedance variations and to calculate the required supply rail voltage and quiescent current accordingly. From feedback I have received, higher quiescent currents tend to sound better so you may wish to bias the compromise between voltage and current accordingly (whilst keeping the power dissipation in the output transistors at a safe level).

 

The following table indicates the maximum power output into 8, 6 and 4ohm loads for some standard transformer secondary voltages, assuming a resistive load and without any allowance for the impedance variations mentioned above. The supply rail voltages assume a regulated supply, with the consequential volt drop, and the quiescent current has been calculated from either the maximum current into 4ohm or, in the case of the 25 and 30Vrms secondary, the transistor power dissipation limit.

 

Secondary

Voltage (Vrms)

Supply Rail

Voltage (V)

Quiescent

Current (A)

Power

8ohm (W)

Power

6ohm (W)

Power

4ohm (W)

18

18

2.8

16

21

32

22

23

3.7

28

37

56

25

28

3.2

42

56

42

30

33

2.7

60

45

30

 

3.  http://www.tinholt.eu/jlhevolution.htm 이것은 아직 분석을 안했다.

Schematic amp
4. 그런데 정말 재미있는 분석은 정작 pass 가 만든 부분이다.   (내용은 첨부 화일 참조)

The PLH Amplifier By Nelson Pass
Introduction: The JLH Amplifier
In 1969 John Linsley쵩ood wrote in Wireless World:
During the past few years a number of excellent designs have been published for
domestic audio amplifiers. However, some of these designs are now rendered
obsolescent by changes in the availability of components, and others are intended to
provide levels of power output which are in excess of the requirements of a normal living
room. Also, most designs have tended to be rather complex.
In the circumstances it seemed worth while to consider just how simple a design could be
made which would give adequate output power together with a standard of performance
which was beyond reproach, and this study has resulted in the present design.
He then described a Class A power amplifier using three gain stages of Bipolar transistors in a
topology which continues to be admired for its elegant simplicity and sound quality.
The centerpiece of this design is the middle stage, an NPN transistor used as a phase splitter,
simultaneously driving the positive half of the output stage and the negative half with symmetric
signals of opposite phase.


Figure 1 shows a simplified version of the JLH topology. Signal input appears at the Base of
Q1, and is amplified and inverted to drive the Base of Q2. Q2 acts as a gain device and also a
signal splitter, driving both Q3 and Q4 simultaneously, but out of phase with each other. Q3 and
Q4 form the output transistors, Q3 operating as a Common Emitter gain device, contributing
current and voltage gain, and Q4 operating as a Common Collector device contributing only
current gain. The resistors provide bias for the system, and R1 and R2 feed the output of the
amplifier in a loop back to the emitter of Q1.
Q2 is the heart of the design, and in my opinion, it is the elegant economy with which it performs
the complementary gain to drive the output devices that gives the circuit its classic beauty.

The JLH was designed at a time when ``the tube era was in decline'' and the new generation of
designers were pulling out all the stops to create big science amplifiers ?pure voltage sources
with high power and infinitesimal distortion -- complex circuits with lots of feedback.
36 years and a little progress later, we can perhaps appreciate the simple charm of the JLH
topology as an exercise in minimalism, but if you haven't listened to one, you might be very
surprised by the quality of sound, which is extraordinarily good within it's power limitations. If
you have efficient speakers and you like to listen to two춃hannel sound at reasonable levels, the
JLH is still in the top rank.
The amplifier has reasonable specifications; nothing special that isn't wildly exceeded by a $3
chip, but it produces real music. Its flaws are not irritating and it does a wonderful job wringing
more music out of modern recordings and even MP3's. I can't think of another transistor design
from that era that works as well.
Figure 2 shows the circuit rendered more completely, but for more extensive documentation on
the versions of the JLH amplifier, I recommend The Class A Amplifier Site:
www.tcaas.btinternet.co.uk
In Figure 2 additional details of setting up the DC bias for each device are shown, where
capacitors are used to separate DC values from AC values. C1 separates feedback from bias
current. C2 separates input signal from input DC bias voltage and C3 blocks the output DC of
the amplifier from the load. C4 removes supply noise from the voltage powering the front end of
the amplifier, and C5 forms a ``bootstrap'' circuit, making resistors R5 and R6 behave more like a
constant current source at audio frequencies.
The original JLH amplifier has approximately 55 dB of open loop gain divided into 22 dB of
amplifier gain and about 33 dB of feedback. As detailed in the original article, it delivered 10
watts at approximately .1% harmonic distortion or less.

The amplifier's popular longevity speaks volumes about the quality of its sound, and this is
understandable given its simplicity coupled with excellent measured performance. It has a
particularly tube춍ike quality compared to the more complex solid춖tate designs of the era and
since.
The distortion is largely 2 nd harmonic, and is closely proportional to the output voltage. This
means that .01% distortion at .1 watts becomes 1% at 10 watts, and you can draw a pretty
straight line between the two points on a logarithmic graph. Such a curve is characteristic of a
single춅nded output topology, and there have been arguments regarding whether or not the
output stage is single춅nded Class A, push춑ull Class A or a mixture of both. We will be having
some fun with that later.
One flaw in the original JLH design was that its bias current, that idling current which flows
through the parts of the circuit, had some dependency on the power supply voltage, resulting in
altered performance for different AC line voltages. Power supply regulation solves this problem
neatly, but there were other ways this was addressed in later versions of the circuit.
Newer JLH Circuits
John Linsley쵩ood published an update to the amplifier in 1996 that addressed bias stability
issues, parts substitutions, and provided a version that had a direct춃oupled output, eliminating
the output capacitor. At the same time, it was in many ways the same amplifier, the measured
performance being very similar.
The JLH circuit continues to be interesting to the audiophile community and has been the
subject of several updates. Figures 3 and 4 show simplified schematics of later generations of
JLH amplifiers.

Figure 3 shows a simplified schematic of the 1996 version published by John Linsley쵩ood that
fixes the bias stability issue with the addition of Z1 and the Q5 portion of the circuit. This version
also direct coupled the output of the amplifier, using dual supply rails.
In 2000 someone else produced the circuit seen in Figure 4, where constant current sources are
used to bias the first two gain stages, giving good power supply rejection to the circuit. This
version also doubled up on the number of output devices. You will notice that Fig 1 -- 4 have
their feedback loop addressing the Emitter of the feedback transistor. Nowadays they have
gotten fancy and call it ``current feedback''.
Just for your entertainment, I cobbled together the circuit of Figure 5 shows an example with a
differential input. An obvious variation, but I haven't seen it used. You can drive this input stage
with a balanced signal by lifting C1 from ground and driving it as a negative input. At 100 ohms
each, R6 and R7 will give this input about the same open loop gain as the original input
transistor with the 220 ohms degeneration of the original feedback impedance.

I recently measured the performance of a working copy of the circuit of Figure 4. It had 17.5 volt
supply rails and was biased at about 2 amps per channel. The open loop gain is also about 55
dB into 8 ohms, and its measured performance is comparable to the original circuit.
Figure 6 shows the distortion versus output power. The bandwidth of the amplifier is --3 dB at
100 KHz, the damping factor is about 35, and the distortion versus frequency is fairly flat, rising
slightly at 20 KHz.
The PLH Amplifier
One of the issues that arise from adding gain stages to amplifiers is that while they increase the
open loop gain and allow more feedback correction, they themselves are the source of
additional distortion. While the extra feedback can lower the distortion numbers, usually the
additional circuitry is reflected in a more complex distortion character having higher order
harmonics and inter춎odulation components. These are generally agreed to be less musical
sounding.
Michael Cunningham wrote, ``Novelists must usually decide what degree of slavish accuracy
would make their stories more alive, and what degree would make them less.'' The amplifier
designer has a similar problem to solve. It's not hard to make an amplifier that measures well --
it's comparatively hard to please audiophiles.
My own approach is to make the signal path as simple as possible, work to lower the distortion
of that basic circuit before feedback is applied, and then apply minimal (or no) feedback, largely
in agreement with the comments in Linsley쵩ood's original article. The result is not always the
best objective measurements, but the sound is often interesting.
The 3춖tage topology of the JLH amplifier routinely uses simple Class A operation and about 33
DB negative feedback to achieve this performance, and this lured me to consider what kind of
amplifier I could achieve with an even simpler circuit and less feedback. The output stage and
the intermediate phase splitter cannot be dispensed with and still resemble a JLH, but you can
certainly remove the input transistor.

4.  diyaudio 에 나오는 쓰레드 2003년의 글에는 여러가지 대화가 나오고 170여 페이지 뷰를 넘어간다.
http://www.diyaudio.com/forums/showthread.php?s=&threadid=3075&perpage=10&pagenumber=36


http://www.diyaudio.com/forums/showthread/t-40355.html



그중의 하나는 없어진 웹사이트에 있던 내용이다.
Graham Maynard presented a jlh output class A amp in the september Electronics World

I뭭e abstracted the input as an ideal gm vccs and used ideal current sources to look at the output stage operation ?

http://www.zero-distortion.com/test...wertrans_05.htm

suggests 2sc3281 is similar to Graham뭩 2sc5200 output devices, I was able to find onsemi spice model for mj3281 and already had fairchild bd139 model for a driver



(despite the LTSwCad file header names this is the mj3281 sim)

my question/issue is the poor current division as Vce gets within 5 V of the rail with the 200 V device, the ancient 2n3055 actually looks much better in this regard (i1 has to go up to ~150mA due to the low hfe of the 3055 model in LtSwCad) ?so is this real or modeling error?

Given that spice transistors are perfectly matched and only deviate due to differential biasing this is probably optimistically good current division so another question is how can the jlh deserve its reputation when built by diyers lacking power curve tracers to match the output devices?

And finally the sim offers an example of how to step a parameter in LtSwCad Spice for mikes

Added spice directive:

.step param A LIST .17 .35 .7 1.4

V5 input voltage source amplitude defined with {A} as parameter which is stepped from LIST in .step directive

SINE(0 {A} 2K)

see LTSpice asc file:

신고
Trackback 2 Comment 2
2010.01.13 09:18

John Linsley Hood 의 Simple Class A Amplifier 번역(JLH AMP)


(아직 완성본이 아니며 임시로 오픈 시켜 몇분에게 피드백을 받기 위한 내용입니다.)
(현재 그럭저럭 보아줄 만한  수준으로 진행중입니다.)
(제작기록과 다른 자료들을 준비하고 있습니다.  jlh는 단순한 주제가 아니랍니다.)
(현재 버전 0 .1 정도 되는 수준입니다. 1.13 )

Simple Class A Amplifier

간단한 클래스 A 앰프

클래스 B보다 주관적으로 좋은 소리를 내는 10와트 디자인

A 10-W Design giving subjectively better results than class B transistor amplifiers

by J. L. Linsley Hood, M.I.E.E.

-역자주-

2010년 오디오 아마추어 안윤호가 번역하고 주석을 달았다. 

번역의 이유는 중요하기 때문이고 사람들이 이 앰프를 만들고 생각하는 일이 중요하다고 생각하기 때문이다. 40년이 넘은 회로지만 이 앰프의 음질은 전설적이어서 아직도 많은 사람들이 만들어보거나 들어보려고 한다. 우리나라에서는 그다지 많이 만들어지지 않았다. 비슷한 디자인의 헤드폰 앰프는 HAS 등의 사이트에서 만들어진 것으로 안다.
그러나 소리를 좋아하는 사람들 , 특히 DIY 성향이 강한 사람들이 이 앰프를 들을 이유는 많다.  

나는 틈나는 대로 이 앰프와 몇 개의 중요한 앰프를 취미의 영역으로 이끌어내어 사람들과 함께 즐겨보고자 한다. PCB도 만들 예정이고 몇 개의 제작사례도 도출할 예정이다. 그리고 JLH만이 아니라 다른 앰프들도 만들고 생각하던 것들을 공유할 예정이다. 히라가도 있고 패스의 알레프나 젠도 있으며 여러가지 다른 장난감들도 있다. 일부는 신비를 떠나 진지한 토이로 돌아올 것이다. 

실제로 나는  2-3년동안 나름대로 많은  앰프를 만들어서 들었으며 어떤 부분은 넬슨 패스(Nelson Pass)와 상의한적도 있다. 친절한 답변에 감사한다.  

아무튼 이제 그 나름대로 아는 것을 공유해야 할 때다. 

첫번째 글은 JLH 에 대한 것이며 몇개 더 있다.  맨처음 글은 번역물로 시작해야 할 것 같다. 

이 글은 1969년 와이어리스월드라는 영국 잡지에 실렸는데 린즐리후드는 유명해졌고 앰프회로를 만들때 항상 약방의 감초처럼 등장하는 구성으로 실려있다. 앞서 말한 넬슨 패스 역시 자신의 버전 PLH (Pass Linsley Hod)를 만들었는데 이때는 2005년이다. 패스는 이 앰프의 소리를 10와트에 만족하기만 한다면 지금도 최고수준이라고 극찬했다.  1969년 이후 몇 개의 변형이 만들어 졌는데 기본적인 회로는 변하지 않았고 소리의 특성도 마찬가지로 모두 JLH 특유의 소리를 갖고 있다. 

당시는 진공관에서 트랜지스터 앰프로 이동하는 시기로 앰프들의 출력은 매우 컸으며 복잡한 회로들이 마구 등장하던 시절이었다.  

이글에 실린 내용은 맥빠질만큼 당연한 내용도 있고 오래된 시절의 지식도 있다. 그러나 어떤 위대한 앰프의 원문이란느 점에서 중요하게 생각한다. 1970년 당시 일본 앰프회로만 주로 공부하던 시절 이런 몇가지 결정적인 회로들을 우리가 열심히 연구했다면 하는 생각이 들기도 한다. 여기에 쓰인 트랜지스터와 비슷한 규격의 2n3055는 그 당시에도 저렴한 트랜지스터에 속했다. 

개인적으로는 앞으로도 사람들이 결정적인 앰프의 소리를 즐기는 것을 권장할 것이고 필요하면 질문도 받고 PCB 같은 것도 공개할 것이다. 좋아할 사람들은 정해져 있겠고 아주 오랜 세월동안 꾸준히 진행할 중요한 게임이다. 

이번의 글은 그 첫발이다. 꾸준히 지치지 않고 진행할 수 밖에 없겠다. 

린즐리후드의 글은 다음과 같이 시작한다:
-------------------------------------------

몇 년동안 가정용 오디오 앰프에 대한 뛰어난 설계들이 발표되었다. 그러나 이들중 일부는 부품의 변경으로 인한 수급문제등으로 폐기되었다. 그리고 다른 것들 중에는 일상적인 거실에 사용하기에는 과한 출력을 내는 것들도 있었다. 또한 대부분의 설계들이 복잡하게 되는 경향이 있었다. 

During the past few years a number of excellent designs have been published for domestic audio amplifiers. However, some of these designs are now rendered obsolescent by changes in the availability of components, and others intended to provide levels of power output which are in excess of the requirements of a normal living room. Also, most designs have tended to be rather complex.

이런 상황에서  적절한 출력과 비난의 대상이 되지 않을 정도의 성능을 내는 아주 단순한 디자인은 가치가 있을 것 같다. 그리고 이 연구의 결과가 현재의 디자인이다. 

In the circumstances it seemed worth while to consider just how simple a design could be made which would give adequate output power together with a standard of performance which was beyond reproach, and this study has resulted in the present design.

출력과 왜율
Output power and distortion

Mullard "5-10" valve amplifier (진공관 앰프)가 아주 흔하다는 것을 생각하면 10와트의 출력은 일상적인 용도에는 충분하게 보인다.  그리고 이들이 스테레오로 사용되고 적절히 효율적인 스피커에 물리면  음향출력은 상당히 클 것이다.
(요즘도 이렇게 생각하는 사람들은 적을 것이나 10와트의 출력이 결코 작지는 않다. 역자주)

In view of the enormous popularity of the Mullard "5-10" valve amplifier, it appeared that a 10-watt output would be adequate for normal use; indeed when two such amplifiers are used as a stereo pair, the total sound output at full power can be quite astonishing using reasonably sensitive speakers.


1947 과 1949에 걸쳐 Wireless World 잡지에 D. T. Williamson 이 제시한 왜율의 기준은  최고출력에서 0.1 % 이하의 값으로 고음질의 오디오 앰프의 목표처럼 되었다.  진공관 앰프와 마찬가지로 트랜지스터 앰프도 트랜스포머를 사용하여 적당한 성능을 얻는 일은 어려운 것이었으나 요즘 트랜지스터 회로는 트랜스포머를 사용하지 않아도 되기 때문에 30hz-20khz 대역에서 최고 출력으로도 0.05% 정도의 왜율이라는 기준을 달성할 수 있을 것으로 보이며 이 구간의 주파수 출력은 균일하다는 것을 의미한다. 


The original harmonic distortion standards for audio were laid down by D. T. Williamson in a series of articles published in Wireless World in 1947 and 1949; and the standard, proposed by him, for less than 0.1% total harmonic distortion at full rated power output, has been generally accepted as the target figure for high-quality audio power amplifiers. Since the main problem in the design of valve audio amplifiers lies in the difficulty in obtaining adequate performance from the output transformer, and since modern transistor circuit techniques allow the design of power amplifiers without output transformers, it seemed feasible to aim at a somewhat higher standard, 0.05% total harmonic distortion at full output power over the range 30Hz-20kHz. This also implies that the output power will be constant over this frequency range.

회로의 설계
Circuit design

윌리엄슨 앰프의 기준에 근접하는 트랜스 없는 디자인의 앰프는 1961년 Tobey and Dinsdale 이 와이어리스 월드에 발표한 디자인다. 이것은 클래스 B 출력단으로 트랜지스터를 의사상보(quasi-complementary) 대칭으로 만들어 놓은 것이다. 그 다음의 고음질 트랜지스터 파워 앰프는 이 기사에 정리한 설계원리를 따르는 경향이 있었다. 

The first amplifier circuit of which the author is aware in which a transformerless transistor design was used to give a standard of performance approaching that of the "Williamson" amplifier, was that published in Wireless World in 1961 by Tobey and Dinsdale. This employed a class B output stage, with a series connected transistors in quasi-complementary symmetry. Subsequent high-quality transistor power amplifiers have largely tended to follow the design principles outlined in this article.

이런 종류 앰프의 큰 장점은 정적인 상태의 전력소모가 아주 낮으며 전반적인 전력변환 효율이 높단느 것이다. 불행히도 태생적인 단점이 있는데 그것은 푸시풀 쌍의 반응이 각기 다르다는 점이다.  또한 Ic/Vb 특성에 따른 크로스오버 왜곡이 존재한다는 점도 있다.  특히 Bailey 같은 사람들이 이런 점들을 시정하기 위해 많은 일을 했다.

The major advantage of amplifiers of this type is that the normal static power dissipation is very low, and the overall power-conversion efficiency is high. Unfortunately there are also some inherent disadvantages due to the intrinsic dissimilarity in the response of the two halves of the push pull pair (if complementary transistors are used in asymmetrical circuit arrangement) together with some cross-over distortion due to the I c /V b characteristics. Much has been done, particularly by Bailey, to minimise the latter.


클래스 B 출력단의 다른 특징은 출력신호가 증가하면 트랜지스터의 전류요구량이 증가하여 설계가 잘 되지 않으면 출력전압이 저하하거나 파워서플라이의 평활이 떨어지게 된다. 또한 출력이 증가함과 동시에 전류가 증가하여 트랜지스터에 대한 일시적인 증가는 열폭주(thermal runaway)로 이어지게 된다. 특히 적절한 보호회로가 없으면 eactive load에서 이런 일이 흔하다. (이 당시는 요즘 secondary break down 이라는 현상을 잘 이해하지 못할 때였다.) 이런 요구사항들이 회로의 구성을 복잡하게 만들고 잘 설계된 클래스 B 앰프들을 더 이상 저렴하거나 간단하게 구성되지 못하게 한다. 

An additional characteristic of the class B output stage is that the current demand of the output transistors increases with the output signal, and this may reduce the output voltage and worsen the smoothing of the power supply, unless this is well designed. Also, because of the increase in current drive with output power, it is possible for a transient overload to drive the output transistors into a condition of thermal runaway, particularly with reactive loads, unless suitable protective circuitry is employed. These requirements have combined to increase the complexity of the circuit arrangement, and a well designed low- distortion class B power amplifier is no longer a simple or inexpensive thing to construct.

*  The thermal runaway referred to is now known to be secondary breakdown, where the transistor suffers from a localised heating on the silicon die. This effect is very rapid, and can lead to almost instantaneous destruction of a transistor. This is one reason that MOSFETs are preferred by many amplifiers designers (the editor is not one of these!).

좋은 성능과 간단한 구성을 갖는 파워앰프에 대한 다른 접근법은 클래스 A 구성으로 출력 트랜지스터를 사용하는 것이다. 이 접근법은 의사 상보형(quasi- complementary) 의 회로의 문제와 트랜지스터의 열폭주 그리고 크로스오버 왜곡과 신호의 크기에 따른 전력 요구량의 문제를 해결한다. 그러나 클래스 B보다 효율이 떨어지며 출력 트랜지스터들은 커다란 방열판에 붙여야 한다.

An alternative approach to the design of a transistor power amplifier combining good performance with simple construction is to use the output transistors in a class A configuration. This avoids the problems of asymmetry in quasi- complementary circuitry, thermal runaway on transient overload, crossover distortion and signal-dependent variations in power supply current demand. It is, however less efficient than a class B circuit, and the output transistors must be mounted on large heat sinks.

기본적인 클래스 A 구성은 단일 트랜지스터를 콜렉터 부하와 함께 구성하는 것으로 그림 1(a)에 나온 것 처럼 저항을 사용하는 것이며  실용적인 해법처럼 보인다. 하지만 전력 변환 효율은 약 12% 정도다. 그림 1(b)에 나오는 것처엄 초크를 사용하면  효율은 증가하지만 가격이 비싸지며 초크의 부피도 커져서 트랜스를 사용하지 않는 설계의 장점이 없어진다.  비슷한 트랜지스터를 콜렉터부하로 사용하는 방법이 그림 1(c)에 나오는데 크기와 가격면에서 편리하며 푸시풀처럼 두번째 트랜지스터를 적당히 반대의 위상으로 구동하여 효율적인 콜렉터 부하로 사용할 수 있다.
이 구성은 트랜지스터를 그림2처럼 구성하여 얻을 수 있다. 

The basic class A construction consists of a single transistor with a suitable collector load. the use of a resistor, as in Fig 1(a), would be a practical solution, but the best power-conversion efficiency would be about 12%. An l.F. choke, as shown in Fig1(b), would give much better efficiency, but a properly designed component would be bulky and expensive, and remove many of the advantages of a transformerless design. The use of a second, similar, transistor as a collector load, as shown in Fig 1(c), would be more convenient in terms of size and cost, and would allow the load to be driven effectively in push-pull if the inputs of the two transistors were of suitable magnitude and opposite in phase. This requirement can be achieved if the driver transistor is connected as shown in Fig. 2.








이 방법으로 연결하는 것은 낮은 왜율 앰프의 가장 중요한 요구사항을 만족시키는 것이다. 그것은 앰프의 근본적인 선형성이 좋아야 한다는 것이며 피드백이 없을 때에도 선형성은 좋아야 한다. 여러 가지 요인이 영향을 미친다. 출력 트랜지스터의 Ic/Vb 비선형성이 없어진다. 한 트랜지스터가 거의 컷오프 상태에 들어갈 때 다른 트랜지스터는 거의 완전히 온 상태에 들어간다. Tr1, Tr2, Tr3에서는 피드백 루프가 만들어지는 방법이 있다.  Tr1 의 베이스 임피던스 특성이 Tr3의 출력 전류에 영향을 미치기 때문이다. 또한 드라이버 트랜지스터 Tr3 (커다란 전압스윙을 가져야 하는데 )은 낮은 하모닉 디스토션에 유리한 조건에서 동작한다. 낮은 출력임피던스와 높은 입력 임피던스다. 이런 구조의 출력단을 갖는 실제회로가 그림3 이다. 

This method of connection also meets one of the most important requirements of a low distortion amplifier :- that the basic linearity of the amplifier should be good, even in the absence of feedback. Several factors contribute to this. There is the tendency of the Ic / Vb non-linearity of the characteristics of the output transistors to cancel, because during the part of the cycle in which one transistor is approaching cut-off the other is turned full on. There is a measure of internal feedback around the loop Tr1 Tr2 Tr3 because of the effect which the base impedance characteristics of Tr1 have on the output current of Tr3. Also, the driver transistor Tr3, which has to deliver a large voltage swing, is operated under conditions which favour low harmonic distortion :- low output load impedance, high input impedance. A practical power amplifier circuit using this type of output stage is shown in Fig. 3.

ESP 사운드의 로드엘리엇은 Tr1의 에미터에 0.1 옴 정도를 달아 전류 피드백을 거는 제안을 하고 있다. 디스토션이 많이 준다는 것이다. 0.1옴은 5와트 이상의 wire wound 저항이다. 





이 회로의 오픈루프 게인은 전형적인 트랜지스터에서 약 600 배정도다.  클로즈드 루프게인은 C3의 임피던스가 R4 에 비해 충분히 낮은 경우 (R3 +R4)/R4 로 정해진다. 그림 3에 나온 값을 따른다면 13이며 피드백 팩터로는 160 밀리 옴에 해당한다. DC에서 이 앰프의 게인은 1이 된다. 왜냐면 C3가 포함된 피드백 루프에서 출력전압 Ve는 Tr4의 베이스 전압에 Tr4의 에미터 전압을 ,   R3에서 일어난 전압강하값을 더한 값과 같기 때문이다. 출력 트랜지스터 Tr1은 Ve를 이 정도 전압까지 끌어 내리기 위해 전류가 필요하며 R1과 R2는 Tr2의 콜렉터 전류를 조절하며 앰프의 정적상태의 전류를 정하는데 사용할 수 있다.  또한 R6과 R5를 조절하여 Ve 값은 원하는 값으로 정할 수 있다.  최적의 성능은 공급전압의 반에 해당하는 값이다. (0.5볼트나 대략 그 정도의 전압정도는 최고출력에 약간의 영향을 미친다, 다른 특성에도 영향은 별로 없다. 이 값이 아주 정확할 필요는 없다.  )


The open loop gain of the circuit is approximately 600 with typical transistors. The closed loop gain is determined, at frequencies high enough for the impedance of C3 to be small in comparison to R4, by the ratio (R3 +R4)/R4. With the values indicated in Fig. 3, this is 13. This gives a feedback factor of about 160 milliohms.
Since the circuit has unity gain at D.C., because of the inclusion of C3 in the feedback loop, the output voltage Ve, is held at the same potential as the base of Tr4 plus the base emitter potential of Tr4 and the potential drop along R3 due to the emitter current of this transistor. Since the output transistor Tr1 will turn on as much current as is necessary to pull Ve down to this value, The resistor R2, which together with R1 controls the collector current of Tr2, can be used to set the static current of the amplifier output stages. It will also be apparent that Ve can be set to any desired value by small adjustments to R5 and R6. The optimum performance will be obtained when this is equal to half the supply voltage. (half a volt or so either way will make only a small difference to the maximum output power obtainable, and to the other characteristics of this amplifier, so there is no need for great precision in setting this.)


사용되는 실리콘 플래너 트랜지스터가  열안정성이 뛰어나고 노이즈 레벨도 적다.  완벽한 대칭성이 필요하지 않으므로 출력단은  실리콘 npn 트랜지스터로 충분하며 성능도 좋고 가격도 저렴하다.  출력부의 성능은 약 10 와트 정도로 윌리엄슨 앰프의 기준을 충분히 충족시킨다. 출력과 게인/주파수 그래프를 그림 4와 5에 보이고 있다. 출력과 전체 하모닉 디스토션의 관계가 그림 6에 보인다. 이 앰프는 그야말로 클래스 A 회로이기 때문에 디스토션은 출력전압에 대해 선형적으로 감소한다. (클래스 B 시스템에서는 상당한 양의 크로스오버 디스토션이 증가하는 경우 반드시 이렇게 되지는 않는다.)  0.05% 정도의 디스토션 요소들을 검사하는 것은 어려운 일이지만 클립핑이 일어나는 지점 이하의  디스토션은 주로 2배수 하모닉이다. 

Silicon planar transistors are used throughout, and this gives good thermal stability and a low noise level. Also, since there is no requirement for complementary symmetry, all the power stages can use n-p-n transistors which offer, in silicon, the best performance and lowest cost. The overall performance at an output level of 10 watts, or at any lower level, more than meets the standards laid down by Williamson. The power output and gain/frequency graphs are shown in Figs. 4 and 5, and the relationship between output power and total harmonic distortion is shown in Fig. 6. Since the amplifier is a straight-forward class A circuit, the distortion decreases linearly with output voltage. (This would not necessarily be the case in a class B system if any significant amount of cross- over distortion was present.) The analysis of distortion components at levels of order of 0.05% is difficult, but it appears that the residual distortion below the level at which clipping begins is predominantly second harmonic.

실리콘 플래나 NPN트랜지스터는 일반적으로 고주파 특성이 뛰어나서 리액티브 부하에 대한 우수한 안정성을 부여한다. 저자는 L 과 C 의 조합으로 시스템이 불안해지는 것을 발견하지 못했다. 그러나 만약 R3이 작은 캐패시터에 의해 션트되어 고주파에서 롤 오프되면 인덕티브(유도)부하에서 쉽게 발진할 수 있다.  


Silicon planar NPN transistors have in general, excellent high frequency characteristics, and these contribute to the very good stability of the amplifier with reactive loads. The author has not yet found a combination of L and C which makes the system unstable, although the system will readily become oscillatory with an inductive load if R3 is shunted by a small capacitor to cause roll-off at high frequencies.


그림 3에 보인 회로는 3-15 오옴에 이르는 부하 임피던스를 극히 작은 변경만으로 구동할 수 있다. 선정된 출력은 각각의 전류/전압의 관계 , 출력트랜지스터의 전류 그리고 출력전압의 스윙을 보이고 있다. 이들은 모두 다르기 때문이다. 피크 전압 스윙과 평균 출력 전류는 간단한 관계 W=I2R 과 V=IR 에 의해 산정할 수 있다. (반드시 기억해야 할 것은 출력은 전압과 전류의 RMS 값으로 계산한다는 것이다. 이들은 피크값을 계산하기 위해 1.41배를 곱해주어야 한다. 그리고 전압 스윙은 피크-피크(첨두값)으로 이것은 피크값의 두배가 된다.)

  


The circuit shown in Fig. 3 may be used, with very little modification to the component values, to drive load impedances in the range 3-15 ohms. However, the chosen output power is represented by a different current/voltage relationship in each case, and the current through the output transistors and the output voltage swing will therefore be different. The peak-voltage swing and mean output current can be calculated quite simply from the well-known relationship W=I2R and V=IR, where the symbols have their customary significance. (it should be remembered, however, that the calculation of output power is based on RMS values of current and voltage, that these must be multiplied by 1.41 to obtain peak values, and that the voltage swing measured is the peak to peak voltage, which is twice the peak value.)



  

 








이 계산들에 의하면 16오옴 로드에 10와트를 만들기 위한 피크-피크 갑슨 34.8 볼트이다. 트랜지스터의 최하 전압이 0.6 볼트이므로 전원은 최하 36볼트를 공급할 수 있어야 gkse. 8 오옴과 3오옴의 부하에서는 27볼트와 17볼트가 필요하다. 그리고 필요한 최저전류는 각각 0.9, 1.2 , 2.0 암페어다. 이들을 테이블 1에 정리했다. C1과 C3은 낮은 주파수에서의 전압과 전원의 롤 오프를 결정하며 그림 4와 5에 나온 값보다 커야 한다.


When these calculations have been made, the peak-to-peak voltage swing for 10 watts power into a 15-Ohm load is found to be 34.8 volts. Since the two output transistors bottom at about 0.

6 volts each, the power supply must provide a minimum of 36 volts in order to su

pply this output. For loads of 8 and 3 ohms, the minimum h.t. line voltage must be 27V and 17V respectively. The necessary minimum currents are 0.9, 1.2 and 2.0 amps. Suggested component values for operation with these load impedances are shown in table 1. C3 and C1 together influence the voltage and power roll-off at low audio frequency performance is desired than that shown in figs. 4 and 5.






공급전압과 출력 전류는 트랜지스터마다 17 와트 정도의 전력소모를 일으키고 부품의 수명을 위해 고온에서 동작하는 것은 좋지 않으니 트랜지스터마다 적절한 방열판을 붙여야 한다,\.   125mm by 100mm (5" by 4") 의 면적에 핀이 붙은 방열판을 트랜지스터마다 붙이는 것을 주장한다. 이것은 어쩔 수 없는 클래스 A가 치러야 하는 대가이다. 30V 가 넘는 공급전압에는 Tr1 과 Tr2 는 Mj481을 Tr3 은 2n1613을 사용해야 한다.



Since the supply voltages and output currents involved lead to dissipation in the order of 17 watts in each output transistor, and since it is undesirable (for component longevity) to permit high operating temperatures, adequate heat sink area must be provided for each transistor. A pair of separately mounted 125mm by 100mm (5" by 4") finned heatsinks is suggested. This is, unfortunately, the penalty which must be paid for class A operation. For supplies above 30V Tr1 and Tr2 should be Mj481s and Tr3 a 2n1613.


만약 프리앰프의 출력임피던스가 수 킬로옴을 넘어간다면 앰프의 입력단은 그림 8처럼 소스 폴로워 방식의 간단한 FET 로 개조할 수 있다. 전체 하모닉 디스토션이 약 0.12 % 정도 증가하므로 좋은 프리앰프를 쓰는 것보다는 덜 매력적인 해결책이다,


If the output impedance of the preamplifier is more than a few thousand ohms, the input stage of the amplifier modified to include a simple f.e.t. source follower circuit shown in fig 8. This increases the harmonic distortion to about 0.12%, and is therefore (theoretically) a less attractive solution than a better pre- amplifier.


만약 높은 주파수의 롤오프가 생긴다면 FET의 게이트와 접지사이에 작은 캐패시터를 연결해 주어야 한다.

A high frequency roll-off can be obtained, if necessary by connecting a small capacitor between the gate of the f.e.t and the negative (earthy) line.


적당한 트랜지스터들 

Suitable transistors


사용하는 트랜지스터들에 따라 회로에서 어떤 차이가 나는 가를 결정하는 실험들을 했다. 예상대로 게인이 높은 트랜지스터에 출력단을 매치해서 사용하면 최고의 성능이 나온다.  드라이버단의 2N697 / 2N1613 에 대해서는 적절한 대체물이 알려진바 없으나 비슷한 타입의 각기 다른 회사의 제품들이 명백하게 동일한 결과를 보였다.  입력 트랜지스터에는 비슷한 규격을 사용하면 차이가 거의 없는 결과를 보였다. Texas Instuments 2N4058은 Motorola 2N3906 와 완벽하게 교체가능하다.


Some experiments were made to determine the extent to which the circuit performance was influenced by the type and current gain of the transistors used. As expected the best performance was obtained when high-gain transistors were used, and when the output stage used a matched pair. No adequate substitution is known for t

he 2N697 / 2N1613 type used in the driver stage, but examples of this transistor type from three different manufacturers where used with apparently identical results. Similarly, the use of alternative types of input transistor produced no apparent performance change, and the Texas Instuments 2N4058 is fully interchangeable with the Motorola 2N3906 used in the prototype.


가장 주목할만한 성능의 차이를 만드는 부분은 출력단 트랜지스터 페어의 전류 게인 특성이다. 가장 적은 디스토션을 원한다면 공급전원의 1/2에서 .25 볼트 이내의 값으로 조절해야 한다. 


The most noteworthy performance changes were found in the current gain characteristics of the output transistor pair, and for the lowest possible distortion with any pair, the voltage at the point from the loudspeaker is fed should be adjusted so that it is within 0.25 volt of half the supply line potential.


실험에 사용한 트랜지스터는 Motorola MJ480/481다 (Texas 2S034를 예외적으로 시도한 적이 있지만 ) 중요한 결론은 트랜지스터의 타입이 중요한 것이 아니라 출력 트랜지스터의 전류게인의 차이가 중요한 것이라는 점이다. 만약 차이가 나는 트랜지스터를 써야 한다면 게인이 높은 것을 Tr1의 위치에 써야 한다. 


The transistors used in these experiments were Motorola MJ480/481, with one exception, in which Texas 2S034 devices were tried. The main conclusion which can be drawn from this is that the type of transistor used may not be very important, but that if there are differences in the current gains of the output transistors, it is necessary that the device with the higher gain shall be used in the position Tr1.



파형의 클리핑이 일어나기전에 디스토션 성분이 나온다면 이들은 거의 2배수의 하모닉이 나타나기 때문이다. 


When the distortion components were found prior to the onset of waveform clipping, these were almost wholly due to the presence of second harmonics.


제작노트 
앰프

Constructional notes
Amplifier

10 + 10 와트 스테레오 앰프의 제작에 필요한 부품들은 'Lektrokit' 이라고 부르는 4 * 4.75 인치의 만능기판에 간편하게 조립할 수 있다. 네 개의 파워 트랜지스터는 외부의 방열판에 붙인다. 특별히 명시하지 않는 한 부품의 값들은 아주 정확할 필요는 없으묘 10% 정도의 오차를 갖는 저항은 아무런 나쁜 효과도 가져오지 않는다.  최소의 잡음레벨은 좋은 부품에서 오며 저항은 카본 필름이나 메탈 옥사이드 저항이 좋다. 

(요즘은 메탈필름 저항이 좋으며 1% 정도의 오차를 갖는다.  카본필름 저항보다 좋다. )


The components necessary for a 10 + 10 watt stereo amplifier pair can be conveniently be assembled on a standard 'Lektrokit' 4" X 4.75" s.r.b.p. pin board, with the four power transistors mounted on external heat sinks. Except where noted the values of components do not appear to be particularly critical, and 10% 
tolerance resistors can certainly be used without ill effect. The lowest noise levels will however be obtained with good quality components, and with carbon-film or metal-oxide resistors.

전원부 :

린즐리후드가  제안한 전원은  capacitance multiplier 라고 부르는 종류다. 
여기에 대해서는 별다른 중요성이 없지만 관심이 있는 사람들은 http://www.tcaas.btinternet.co.uk/index-1.htm 에 있는 글을 읽는 것을 권한다.  다른 문헌은 로드엘리엇의 글(http://sound.westhost.com/project15.htm)을 읽는 것도 좋은 출발점이 될 것이다. 



- 완역본은 아니지만 원본의 대강은 이 정도면 충분히 파악할 수 있다. 1969년 진공관에서 트랜지스터로 넘어가던 시절 잡지에 발표하기를 조금 주저하던 린즐리 후드는 이 글로 오디오구루가 된 자신을 발견했다고 한다. 그 후에도 많은 글들을 썼으나 이글이 가장 결정적이다. 


참고로 클래스 A 앰프의 개략적인 구성은 D.Self의 교과서에 따르면 다음과 같은 몇개의 구성밖에는 없다.

그림 D를 제외하면 위상 반전을 하는 tr1의 존재는 아주 특이한 것이다.





로드 엘리엇은 이 글을 소개하면서 린즐리 후드를 이렇게 소개했다,

The Author


저자 

John L Linsley-Hood는 왕성한 활동을 하는 앰프 디자인의 저자이며  Electronics World (과거의 Wireless World)잡지에 여전히 새로운 아이디어와 회로들을 삳고 있다.  고성능 오디오 앰프에 미친 그의 영향은 상당하며 요즘도 계속 이어지고 있다. 이것은 내가 그의 아이디어나 이론을 흠모하거나 동의 하는 것을 의미하는 것은 아니지만 적어도 린즐리후드는 자신이 생각하는 것을 주장하는 용기가 있는 사람이다. 물론 잡지사 역시 그것을 출판해줄 용기가 있다.


John L Linsley-Hood is a prolific author of amplifier designs, and still presents new ideas and circuits in the UK magazine Electronics World (formerly Wireless World). His influence on the design of quality audio amplifers has been considerable, and continues to this day. This is not to say that I agree with or endorse all his ideas or theories, but at least he has the guts to say what he thinks, and the magazine has the guts to print it, too.


역주)  존 리즐리 후드는 2004년 타계했는데 위키피디어에는 다음과 같이 소개하고 있다.

John Linsley Hood was an electronics designer who is best remembered for his "Simple Class A Amplifier", which he designed to provide a good-quality performance comparable with that of the classic Williamson amplifier.

The design was published in Wireless World in 1969 (April 1969 issue, p. 148), and later updated in 1996.

He wrote for a number of magazines and published a number of books including

  • The Art of Linear Electronics (Oxford, Butterworth-Heinemann, 1993);
  • Audio Electronics (Oxford, Newnes, 1995); and
  • Valve and Transistor Audio Amplifiers (Oxford, Newnes, 1997).

He was born in 1925, and died on 11 March 2004.

John Laurence Linsley-Hood, born 1925,was educated at Reading School,Acton Polytechnic,the Royal Technical College(Glasgow)and after war,at Reading University.In 1942 he joined the G.E.C.Research Laboratries at Wembley,working on magnetron development as junior member of a team. In 1943 he joined the R.A.F in aircrew but was transferred to work on Radar. He subsequently worked with T.R.E.(Malvern)overseas. After a return to university he joined the Windscale Research Laboratories of the Atomic Energy Authority He has been in charge of the electronics team in the research Laboratories of British Cellophane Ltd.since 1954.




저작자 표시
신고
Trackback 0 Comment 0


티스토리 툴바